Power-supply device

ABSTRACT

A step-down type DC-DC power-supply device implements both the stabilization of the control loop and the responsibility at the same time. In the power-supply device, an output power signal is fed back to an error amplifier after having passed through a CR smoothing filter provided independently of a power LC smoothing filter. Also, independently of the duty controls over Power MOSFETs, i.e., upper-side/lower-side semiconductor switching components in the steady state, an output from the power LC smoothing filter is added to an upper and lower limit-mode-equipped control circuit, thereby, at the transient state, forcefully setting the duty α at either 0% or 100%.

BACKGROUND OF THE INVENTION

[0001] 1. Field of the Invention

[0002] The present invention relates to a power-supply device where,independently of a power LC smoothing filter, a signal is caused to passthrough a CR smoothing filter and is then fed back so that the controlloop will be stabilized.

[0003] 2. Description of the Related Art

[0004] A prior art on the loop stabilizing method for a power-supplydevice has been described in “Low-Voltage On-Board DC/DC Modules forNext Generations of Data Processing Circuits”, Zhang et al., IEEE Tran.on Power Elect. Vol. 11, No. 2, March 1996. In the power-supply deviceaccording to the prior art, a signal is fed back to an error amplifierfrom a power LC smoothing filter. Then, the error amplifier compensatesthe phase, thereby implementing the stabilization of the control loop.In this prior art, an aluminum electrolytic capacitor is used as thepower LC smoothing filter.

[0005] U.S. Pat. No. 5,877,611 discloses a power supply system in whichan output of a CR smoothing filter connected across an inductor of anoutput LC smoothing filter is fed back to an error amplifier having alow input impedance. In the U.S. patent prior art, voltage and currentsignals of a power supply output are extracted using the CR smoothingfilter, so that the resistance value of the CR smoothing filter must beset to be small. The component constants of the CR smoothing filter area capacitance of 0.47 μF and a resistance of 100 Ω. Accordingly, the CRsmoothing filter having such constants cannot be formed on chip in apower supply IC and must be formed externally of the IC chip, resultingin a problem that the power supply device cannot be made in small sizetotally.

SUMMARY OF THE INVENTION

[0006] In order to downsize the power-supply device, instead of usingthe aluminum electrolytic capacitor as the power LC smoothing filter,there has occurred a necessity for using a ceramic capacitor of achip-part as the power LC smoothing filter. However, the equivalentseries resistance (ESR) of the chip ceramic capacitor is equal toseveral mΩ, which is considerably small. What is more, the ceramiccapacitors are connected in parallel under an actual use condition.Accordingly, the total of the ESRs in this case becomes less than 1 mΩ,which is even smaller. This makes it impossible to expect the damping ofthe ESR as is expected in the case of using the aluminum electrolyticcapacitor. Consequently, it becomes difficult to stabilize the controlloop.

[0007] In the above-described prior art, when using the ceramiccapacitor with the small ESR as the power LC smoothing filter, itbecomes impossible to expect the damping effect of the ESR. This causesa signal to oscillate, thereby making the phase compensation difficult.Also, if, in the prior art, it were to become possible to implement thephase compensation by narrowing the operation bandwidth of the erroramplifier, a response from the power-supply is delayed exceedingly.Moreover, in modifying the LC smoothing filter's constants, there existsa troublesome task of adjusting the phase compensation condition of theerror amplifier on each that occasion.

[0008] It is an object of the present invention to provide apower-supply device that employs a novel control method where,independently of a power LC smoothing filter, a signal is caused to passthrough a CR smoothing filter and is then fed back so that the controlloop will be stabilized.

[0009] A power-supply device according to one aspect of the presentinvention is as follows: In the control loop of the power-supply deviceof a step-down type DC-DC converter, a CR smoothing filter is providedindependently of a power LC smoothing filter. Moreover, a signalcorresponding to the output power is fed back to an error amplifierafter having passed through the CR smoothing filter.

[0010] Also, a power-supply device according to another aspect of thepresent invention includes the following unit: Independently of the dutycontrols over Power MOSFETs, i.e., upper-side/lower-side semiconductorswitching components in the steady state, the unit adds the output froma power LC smoothing filter to an upper and lower limit value detectingcircuit, thereby, at the transient state, forcefully setting the duty ateither 0% or 100%.

[0011] Moreover, a power-supply device according to still another aspectof the present invention is as follows: The power-supply device includespower-supply device units prepared in plural number. In order to performa parallel operation of these power-supply device units, thepower-supply device further includes an oscillator and a phase shiftcircuit that the plural power-supply device units have in common.Furthermore, in the steady state, phases of driving pulses ofupper-side/lower-side Power MOSFETs in the respective power-supplydevice units are respectively shifted to phases that result fromdividing 360° by the number of the parallelism. At the transient state,all of the parallel power-supply device units are operated by drivingpulses of one and the same phase.

[0012] Other objects, features and advantages of the invention willbecome apparent from the following description of the embodiments of theinvention taken in conjunction with the accompanying drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

[0013]FIG. 1 is a circuit block diagram for illustrating a power-sourcedevice of a first embodiment in the present invention;

[0014]FIG. 2 is an explanatory diagram for explaining an IC where a CRfilter is built in a semiconductor chip in the power-supply device inFIG. 1;

[0015]FIG. 3 is a circuit block diagram for illustrating a power-supplydevice of a second embodiment in the present invention;

[0016]FIG. 4 is an explanatory diagram for explaining an IC where a CRfilter is built in a semiconductor chip in the power-supply device inFIG. 3;

[0017]FIG. 5 is a circuit block diagram for illustrating a power-supplydevice of a third embodiment in the present invention;

[0018]FIG. 6 is a circuit diagram for illustrating the details in FIG.5;

[0019]FIG. 7 is a diagram for illustrating the operation state mode inFIG. 6;

[0020]FIG. 8 is a circuit block diagram for illustrating a power-supplydevice of a fourth embodiment in the present invention;

[0021]FIG. 9 is a circuit block diagram for illustrating anotherpower-supply device of the fourth embodiment;

[0022]FIG. 10 is a circuit block diagram for illustrating still anotherpower-supply device of the fourth embodiment;

[0023]FIG. 11 is a circuit block diagram for illustrating a multi-phasepower-source device of a fifth embodiment in the present invention;

[0024]FIG. 12 is a circuit diagram for illustrating the details in FIG.11;

[0025]FIG. 13 is a diagram for illustrating the operation state mode inFIG. 12;

[0026]FIG. 14 is a circuit block diagram for illustrating an example ofthe chip configuration of a power-source device of a sixth embodiment inthe present invention;

[0027]FIG. 15 is an explanatory diagram for explaining a VID code inputD/A converter applied to FIG. 14;

[0028]FIG. 16 is a circuit block diagram for illustrating a multi-phasecompatible chip of a seventh embodiment in the present invention;

[0029]FIG. 17 is an explanatory diagram for explaining the printedwiring board implementation of a power-source control IC of an eighthembodiment;

[0030]FIG. 18 is an explanatory diagram for explaining a HDD device of aninth embodiment;

[0031]FIG. 19 is an explanatory diagram for explaining a tenthembodiment in the present invention;

[0032]FIG. 20 is an explanatory diagram for illustrating anotherembodiment of a pulse-width modulation oscillator PWM;

[0033]FIG. 21 is an explanatory diagram for explaining an eleventhembodiment in the present invention applied to a commercially-availablepower-source IC; and

[0034]FIG. 22 is a diagram for illustrating the operation state mode inFIG. 21.

DETAILED DESCRIPTION OF THE EMBODIMENTS

[0035] Referring to the accompanying drawings, the explanation will begiven below concerning the details of the present invention.

[0036] Embodiment 1

[0037]FIG. 1 illustrates a power-supply device of the presentembodiment. In FIG. 1, reference notations Vi and Vo denote an inputterminal and an output terminal, respectively. An upper-side PowerMOSFET Q1 is connected to the input terminal Vi, and a lower-side PowerMOSFET Q2 is connected to a ground potential side. An LC smoothingfilter, i.e., a power output filter consisting of an inductor L and acapacitor Co, and a CR smoothing filter consisting of a resistor R and acapacitor C are connected in parallel to a midpoint of the Power MOSFETsQ1 and Q2. Moreover, the output terminal Vo is connected to a midpointof the LC smoothing filter, and one input (−) of an error amplifier EAis connected to a midpoint of the CR smoothing filter. Here, thecapacitor Co of the LC smoothing filter is a chip ceramic capacitor.

[0038] Also, a reference voltage Vref is connected to the other input(+) of the error amplifier EA. A pulse-width modulation (abbreviated asPWM) oscillator PWM, and gates of the Power MOSFETs Q1 and Q2 via adriver DRV are connected to an output of the error amplifier EA. ThePower MOSFETs Q1 and Q2 are driven in opposite phases to each other, andthus are electrically conducted alternately. In the present embodiment,an output voltage Vout is smaller than an input voltage Vin.

[0039] Next, the explanation will be given below regarding the circuitoperation in FIG. 1. The input voltage Vin applied to the input terminalVi is converted into a voltage by on/off controls over the upper-sidePower MOSFET Q1 and the lower-side Power MOSFET Q2 via the CR smoothingfilter. This converted voltage VFB is compared with the referencevoltage Vref by the error amplifier EA. As a consequence, an errorvoltage is generated in a state of being amplified at the output of theerror amplifier EA. This error voltage is converted into a PWM pulse bythe pulse-width modulation oscillator PWM. This PWM pulse is furtherconverted by the driver DRV into an on/off-time ratio (i.e., duty: α) atwhich the driver DRV drives the upper-side Power MOSFET Q1 and thelower-side Power MOSFET Q2. Moreover, a negative-feedback control isperformed over the PWM pulse so that the error voltage becomes equal to0. As a result of this, the converted voltage VFB becomes equal to thereference voltage Vref. In this case, the converted voltage VFB acquiredthrough the CR smoothing filter in the steady state is proportional tothe duty α of the input voltage Vin. Consequently, the followingrelational expression holds:

VFB=Vref=α·Vin

[0040] where the duty α assumes a value in the range of 0 to 1, since αis defined as the on-time/(a total of the on-time and the off-time).

[0041] In the case of the ordinary step-down type converter, it has beenfound out that the voltage-converted ratio in the steady state is equalto the ratio, i.e., the duty, between the output voltage and the inputvoltage. Accordingly, assuming that the input voltage is Vin and theduty is α, the output from the LC smoothing filter, i.e., the outputvoltage Vout acquired at the output terminal Vo, can be determined by arelational expression:

Vout=α·Vin.

[0042] From the above-described 2 expressions, the following relationalexpression holds:

Vout=VFB=α·Vin.

[0043] Consequently, even if no direct negative-feedback control isperformed over the output from the LC smoothing filter, if an indirectcontrol over the duty α using some other method proves successful, thissuccessful indirect control becomes equivalent to a direct control overthe output voltage Vout at the output terminal Vo. As a result, itbecomes possible to acquire, at the output terminal Vo, the voltage thatis proportional to the duty α of the input voltage Vin. In other words,the Power MOSFETs Q1 and Q2 are driven, thereby performing thenegative-feedback control over the output from the CR smoothing filter.This operation allows the desired voltage, which is proportional to theduty α of the input voltage Vin, to be also acquired at the output fromthe LC smoothing filter as the output voltage Vout.

[0044] As the voltage converting method based on the duty control overthe upper-side Power MOSFET Q1 and the lower-side Power MOSFET Q2, thepresent embodiment is a primary-delay control method where the CRsmoothing filter is used for the control loop. Accordingly, since thereexists none of the secondary delay by the LC smoothing filter as wasfound in the prior art, the control loop does not become the oscillatingsystem. This prevents the oscillating waveform from occurring in theoutput, thereby making the loop stable. Consequently, according to thepresent embodiment, even if the chip ceramic capacitor with a small ESRis used as the capacitor of the LC smoothing filter, it is possible tostabilize the control loop.

[0045] Next, the explanation will be given below concerning thelarge-or-small relationship among the corner frequencies and theswitching frequency of the above-described 2 smoothing filters. Let'sassume that the corner frequency of the CR smoothing filter and that ofthe LC smoothing filter are equal to fCR and fLC respectively, and thatthe switching frequency is equal to fSW. Then, setting the relationshipamong these frequencies as fLC<fCR<fSW makes it possible to ensure thestability of the loop. Moreover, from this relationship, the feedbackfrom the CR smoothing filter results in a higher operation frequency ascompared with the feedback from the LC smoothing filter, which allowsthe implementation of the high-speed response. Also, fLC and fCR are setas frequencies that are different to some extent. This setting, even ifthe LC smoothing filter's constants are modified, makes it unnecessaryto change the CR smoothing filter's constants, thereby allowing anincrease in the degree-of-freedom of the design. With respect to thehigh-speed operation of a 1-to-6-MHz switching frequency, values usableas the LC smoothing filter's constants and the CR smoothing filter'sconstants are, e.g., 0.2 μH, 220 μF, and 20 μF, 200 kΩ, respectively. Ifthe values of these capacitors and this resistor are of these orders, itbecomes possible to mount (i.e., on-chip) the CR smoothing filter on asemiconductor integrated circuit chip, thereby makingexternally-attached components unnecessary. This means the following: Bymerely replacing the power-supply device illustrated in FIG. 1 by an ICwhose terminal location is the same (i.e., pin-compatible) as that ofthe power-supply control IC in the prior art, the printed wiring boardin the prior art can be utilized with no modification added thereto.

[0046]FIG. 2 is an explanatory diagram for explaining the chip layout inthe case where, in the power-supply device in FIG. 1, the CR smoothingfilter is built in a semiconductor chip. In FIG. 2, reference notationsC and R denote a built-in capacitor and a built-in resistor,respectively. These components are mounted on a semiconductor board thatis the same as the one that mounts thereon the error amplifier EA, thepulse-width modulation oscillator PWM, the driver DRV, and the PowerMOSFETs Q1 and Q2.

[0047] So far, the explanation has been given selecting, as the example,the CR smoothing filter whose output is fed back to the error amplifierin the control loop. Instead of the CR smoothing filter, however, theuse of another high-response filter circuit allows the acquisition ofbasically the same effects. Also, although the explanation has beengiven selecting the Power MOSFETs as the example of the semiconductorswitching components, the IGBTs may be used instead.

[0048] Embodiment 2

[0049]FIG. 3 illustrates the present embodiment. In FIG. 3, the samereference notations are attached to the same configuration components inFIG. 1. The point in which FIG. 3 differs from FIG. 1 is that the CRsmoothing filter is set up at both ends of the inductor L of the LCsmoothing filter. In the present embodiment, since the electrostaticcapacitance of the capacitor Co of the output LC smoothing filter islarge, the inductor-connected edge side of the capacitor Co can also beregarded as the ground potential. The present embodiment also allows theacquisition of basically the same effects in FIG. 1. Furthermore, thepresent embodiment makes it possible to perform the negative feedback ofan infinitesimal capacitance change caused by a temperature change inthe capacitor Co of the LC smoothing filter. Consequently, even if thechip ceramic capacitor with a small ESR is used, the present embodimentpermits an enhancement in the stability of the control loop. In thiscase as well, the constants of the embodiment in FIG. 1 are usable asthe CR smoothing filter's constants. FIG. 4 illustrates an explanatorydiagram for explaining the chip layout in the case where, in thepower-supply device in FIG. 3, the CR smoothing filter is built in asemiconductor chip.

[0050] Embodiment 3

[0051]FIG. 5 illustrates a power-supply device obtained by furtherproviding a transient variation detecting circuit TVD into the 1stembodiment. This transient variation detecting circuit TVD controls theduty of the pulse-width modulation oscillator PWM by detecting atransient load variation between the output voltage Vout at the outputterminal Vo and a voltage that results from adding a upper and lowerlimit-voltage width ±Δ to the reference voltage Vref. FIG. 6 illustratesa concrete example of the pulse-width modulation oscillator PWM and thatof the transient variation detecting circuit TVD.

[0052] In FIG. 6, the pulse-width modulation oscillator PWM is avariable oscillator including a voltage-to-current converting circuitV/I, current-source MOSs 110, 120, inverters INV11, INV12, a capacitor105, and a flip-flop FF. Also, the transient variation detecting circuitTVD includes comparators CMP1, CMP2, switching MOSs SW1 to SW4, constantcurrent-sources I1 to I4, and inverters INV1 to INV8.

[0053] The transient variation detecting circuit TVD includes a windcomparator consisting of the 2 comparators CMP1, CMP2. The circuit TVDcompares the output voltage Vout with the voltage that results fromadding the upper and lower limit-voltage width ±Δ to the referencevoltage Vref, thereby detecting the operation state of the outputvoltage Vout and determining the pulse duty α of the oscillator PWMindicated in FIG. 7. This means that, in the transient variationdetecting circuit TVD, the control method in the steady state and theone at the transient state are switched into control modes that matchthe operation state.

[0054] From the outputs from the 2 comparators CMP1, CMP2, the following3-way information is acquired: (a) a case where the load current isdecreased, (b) the steady state, (c) a case where the load current isincreased. Using FIG. 7, these cases will be explained below:

[0055] The case (a) is under a condition Vout≧(Vref+Δ). At this time,the output duty α of the pulse-width modulation oscillator PWM isforcefully set at 0%. For this purpose, the switching MOSs SW1 and SW4are turned on, and the switching MOSs SW3 and SW2 are turned off. As aresult, a current from the constant current-source I1 is added to acurrent from the current-source MOS 110, then flowing together to theinverter INV11. A current from the constant current-source I4 issubtracted by a current to the current-source MOS 120, so that thecurrent value flowing to the inverter INV12 becomes equal to 0.Consequently, the upper-side Power MOSFET Q1 is switched off, and thelower-side Power MOSFET Q2 is switched on, which, eventually, makes theduty α equal to 0%. In this case, in order to set the duty α at 0%completely, it is preferable that current values from the constantcurrent-sources I1 to I4 be each set at the total current ofdifferential pair operation currents of the voltage-to-currentconverting circuit V/I.

[0056] The case (b) is under a condition (Vref+Δ)>Vout>(Vref−Δ). In thiscase, all of the switching MOSs SW1 to SW4 are turned off, and areoperated in accordance with a current ratio determined by a controlinstruction from the error amplifier EA. Since this current ratio isequal to the rate of the duty, the voltage that is proportional to theduty a of the input voltage Vin can be acquired as the output voltageVout.

[0057] The case (c) is under a condition Vout≦(Vref−Δ), where the duty αis forcefully set at 100%. In this case, the switching MOSs SW3 and SW2are turned on, and the switching MOSs SW1 and SW4 are turned off. As aresult, a current from the constant current-source I3 is added to thecurrent from the current-source MOS 120, then flowing together to theinverter INV12. A current from the constant current-source I2 issubtracted by the current to the current-source MOS 110, so that thecurrent value flowing to the inverter INV11 becomes equal to 0.Consequently, the upper-side Power MOSFET Q1 is switched on, and thelower-side Power MOSFET Q2 is switched off, which, eventually, makes theduty α equal to 100%. In this case, in order to set the duty α at 100%completely, it is preferable that the current values from the constantcurrent-sources I1 to I4 be each set at the total current of thedifferential pair operation currents of the voltage-to-currentconverting circuit V/I.

[0058] The present embodiment forcefully switches the duty α of thepulse-width modulation oscillator PWM to either 0% or 100% so that thevoltage generated at the output terminal Vo at the transient state willfall within the upper and lower limit-voltage width ±Δ added to thereference voltage Vref. This rapidly suppresses the output voltage Voutwithin (Vref±Δ). Moreover, when the operation state enters the steadystate, the present embodiment causes the output voltage to be stabilizedas the voltage that is proportional to the duty α of the input voltage.

[0059] In this way, in the present embodiment, the control mode isautomatically switched depending on whether the operation state is thetransient state or the steady state. As a consequence, with respect toeven, e.g., an about 10A transient load variation having the highcurrent slew rate (i.e., di/dt) of 500A/μs, it becomes possible tosimultaneously implement both the high-speed response and thestabilization of the output voltage in the steady state.

[0060] Next, using FIG. 20, the description will be given belowconcerning another embodiment of the pulse-width modulation oscillatorPWM. A circuit illustrated in FIG. 20 can be implemented by acombination of an oscillator OSC, a one-shot multivibrator OSM, and aV/I converter V/I. A constant time-period pulse can be generated by theoscillator OSC as follows: A MOS 130 and a constant current-source I5set a constant current which is needed for determining the desiredtime-period. Next, this constant current is made to flow to thecurrent-source MOSs 110, 120 of the pulse-width modulation oscillatorPWM in FIG. 6. Also, when this constant time-period pulse is applied toa clock terminal CLK of the one-shot multivibrator OSM, the terminalvoltage of a capacitor CT becomes equal to 0 on a temporary basis. Atthe next moment, however, the capacitor CT is electrically charged by acurrent that results from converting the error voltage of the erroramplifier EA by the V/I converter V/I. Moreover, a time that has elapseduntil this charge voltage attains to a predetermined threshold value isacquired as the PWM pulse. In this way, the series of pulse-widthmodulation oscillating operations can be repeated. Namely, it becomespossible to acquire the PWM pulse that is proportional to the errorvoltage of the error amplifier EA.

[0061] This pulse-width modulation oscillator PWM is used as aneffective unit in a multi-phase control in FIG. 11 and FIG. 12 whichwill be described later. In this case, in order to implement themulti-phase operation, a phase shift circuit needs to be inserted afterthe oscillator OSC.

[0062] Embodiment 4

[0063] FIGS. 8 to 10 illustrate the present embodiment. The embodimentin FIG. 8, which is obtained by providing the transient variationdetecting circuit TVD into the embodiment in FIG. 3, allows theacquisition of basically the same effects in FIG. 5. The configurationsin FIG. 9 and FIG. 10 are as follows: In the circuit diagrams in FIG. 1and FIG. 3, the input into the transient variation detecting circuit TVDis drawn from the midpoint of a series circuit that consists of acapacitor C3 and a resistor R3 which are set up at both ends of theinductor L of the LC smoothing filter. As a result of this, the phase ofan inductor L current, which can be detected by the series circuit ofthe capacitor C3 and the resistor R3, and the charge/discharge phase ofthe output capacitor Co can be made to coincide with each other.Consequently, it becomes possible to eliminate as much as possibleexcessive/redundant electric charges produced by the charge/discharge ofthe output capacitor Co from the inductor L current. This makes itpossible not only to implement the high-speed response and the highstability, but also to reduce a variation (i.e., ripple) in the outputvoltage at the transient state.

[0064] Embodiment 5

[0065] The present embodiment is a multi-phase embodiment where theplural power-supply device units in the 1st to the 4th embodiments areoperated in parallel. The present embodiment combines the 2 or moresame-type power-supply devices indicated in the 1st to the 4thembodiments. Hereinafter, the explanation will be given below selectingthe 2-phasing as the example.

[0066]FIG. 11 illustrates the embodiment that results from multi-phasingthe power-supply device unit in FIG. 8. In order to implement themulti-phasing, the embodiment in FIG. 11 newly includes the oscillatorOSC and a phase shift circuit PSFT, which generate 2-phase pulses whosephases are shifted to each other by 180°. This embodiment inputs each ofthe 2-phase pulses into each of pulse-width modulation oscillators PWM1and PWM2, thereby implementing the multi-phase control.

[0067]FIG. 12 illustrates, in more detail, the embodiment of thepower-supply device in FIG. 11. In FIG. 12, the pulse-width modulationoscillator PWM1 includes a voltage-to-current converting circuit V/I1and a one-shot multivibrator OSM1. In the steady state, the oscillatorPWM1 operates by receiving a pulse signal from the phase shift circuitPSFT.

[0068] Using an operation state mode in FIG. 13, the explanation will begiven below regarding the operation of the embodiment in FIG. 12. Thisoperation state mode will be explained in much the same way as the caseof the 3rd embodiment. Hereinafter, the explanation will be givenconcerning the Phase 1 power-supply illustrated on the upper-half sidein FIG. 12.

[0069] (a) In the case of Vout≧(Vref+Δ), the output duty of thepulse-width modulation oscillator PWM1 is forcefully set at 0%. For thispurpose, the reset RST of the one-shot multivibrator OSM1 is turned on,which makes the duty equal to 0%.

[0070] (b) In the case of (Vref+Δ)>Vout>(Vref−Δ), as an ordinaryoperation of the one-shot multivibrator, the OSM1 receives the pulsefrom the phase shift circuit PSFT as a clock CLK, thereby generating anon-pulse width. The on-pulse width is determined by the current valuefrom the current-source MOS 210 and the capacitance value of a capacitorCT1, i.e., a timing capacitor. This on-pulse width is of a control modethat operates in accordance with the current ratio determined by thecontrol from the error amplifier EA. Namely, since this current ratio isequal to the duty, the output voltage Vout becomes equal to the voltagethat is proportional to the duty α of the input voltage Vin.

[0071] (c) In the case of Vout≦(Vref−Δ), the duty is forcefully set at100%. For this purpose, both ends of the capacitor CT1, i.e., the timingcapacitor, are short-circuited by a MOS switch M21 so as to maintain theon-state, which makes the duty equal to 100%. Incidentally, a detectionresult by an overcurrent detecting circuit OCI is also added to thereset RST, thereby preventing a component breakdown caused by anovercurrent from the upper-side Power MOSFET Q1. Concerning the Phase 2power-supply on the lower-half side in FIG. 12, the explanation will beomitted because the operation is the same as the Phase 1 power-supply.

[0072] In the operations described so far, in the steady state, theinductor currents from the 2 power-sources operate in opposite phases,i.e., in phases shifted to each other by 180°. Meanwhile, at thetransient time, the inductor currents from the 2 power-supplies becomethe same in their phases, thereby dealing with a rapid load variation.The present embodiment not only increases the output current by usingthe plural power-supplies, but also reduces a ripple in the outputvoltage.

[0073] In the case of providing the 2 or more power-supply device units,there are provided an oscillator and a phase shift circuit that theplural power-supply device units have in common. Moreover, in the steadystate, phases of driving pulses of the upper-side/lower-side PowerMOSFETs in the respective power-supply device units are respectivelyshifted to phases that result from dividing 360° by the number of thepower-supply device units located in parallel. At the transient state,as are the cases with the above-described (a) and (c), all of theparallel power-supply device units are operated by driving pulses of oneand the same phase. In the case of, e.g., the 4 power-supply deviceunits, it is advisable to shift the phases to the respective phases of0° (i.e., criterion), 90°, 180°, and 270°.

[0074] Embodiment 6

[0075] Next, the explanation will be given below concerning anembodiment of the IC chip configuration of the power-supply controldevice in the present invention.

[0076]FIG. 14 illustrates the embodiment of the one-chip configurationof the circuit configuration illustrated in FIG. 8. In FIG. 14, circuitsand functions are all implemented on-chip on one semiconductor boardexcept for the following externally-mounted components: The LC smoothingfilter, the CR circuit consisting of the capacitor C3 and the resistorR3 for detecting the current phase of the transient variation detectingcircuit TVD, and a boost circuit consisting of a diode DBT and acapacitor CBT.

[0077] The on-chip implemented circuits and functions are as follows:The CR smoothing filter consisting of the capacitor C and the resistorR, the error amplifier EA, the reference voltage Vref, the pulse-widthmodulation oscillator PWM, a dead band circuit DBU, a dead band circuitDBL, a level shift circuit LS, a driver DRVU, a driver DRVL, theupper-side/lower-side Power MOSFETs Q1, Q2, an overcurrent detectingcircuit OC, the transient variation detecting circuit TVD, an upper andlower limit-voltage generating circuit VΔ, a soft-start circuit SS, anunder-voltage lockout circuit UVLO, and a power-good circuit PWRGD.Incidentally, instead of acquiring the reference voltage Vref from aband-gap reference circuit, the reference voltage Vref may be acquiredby receiving a digital signal corresponding to a VID (: VoltageIdentification) code, using an on-chip D/A converter illustrated in FIG.15. Although there exist not-illustrated circuits and functions, the1-chip power-supply control IC in the present embodiment is equippedwith the functions implemented in compliance with the VRM 9.1 expoundedby the Intel Corporation.

[0078] Although, in FIG. 14, the explanation has been given selectingthe case where the upper-side Power MOSFET Q1 is the NMOS, the MOSFET Q1may also be a PMOS. In this case, the externally-mounted boost circuitbecomes unnecessary. However, since it is necessary to drive the gate ofthe PMOS at the electric potential from the input terminal Vi, avoltage-generating supply for this necessity is implemented on-chip.

[0079] The voltage fed to the input terminal Vi and the one fed to apower-supply terminal Vcc may be made equal to each other, e.g., 5V or12V. Otherwise, the voltages may be made different, e.g., 12V is fed tothe input terminal Vi, and 5V is fed to the power-supply terminal Vcc.When the voltage fed to the input terminal Vi and the one fed to thepower-supply terminal Vcc are different, 5V to the power-supply terminalVcc may be fed from the outside. Otherwise, 5V may be generated by theon-chip circuit from 12V fed to the input terminal Vi, then beingsupplied thereto. Incidentally, when feeding 12V to the input terminalVi, an about 7V Zener diode is connected to the boost circuit in FIG. 14in series with the diode DBT, thereby preventing the gate voltage of theupper-side Power MOSFET from becoming too large.

[0080] Also, in the operation of the soft-start circuit, at the time ofinjecting the power-supply, it is preferable to mask the output signalfrom the transient variation detecting circuit for the high-speedresponse.

[0081] Embodiment 7

[0082]FIG. 16 illustrates a multi-phase-compatible IC chip configurationin the present embodiment. The configuration in FIG. 16 results frommulti-phasing the circuit configuration of the IC chip illustrated inFIG. 14. The point that differs from the 6th embodiment in FIG. 14 isthat the oscillator OSC and the phase shift circuit PSFT are added tothe IC chip. As IC pins that become necessary for implementing themulti-phasing, there exist terminals for providing its-own/the other ICchips with phase pulses φ1 to φ4 that correspond to the number of themulti phases, and terminals for supplying the reference voltage Vref,and outputs from the upper and lower limit-voltage generating circuit VΔto the transient variation detecting circuit TVD.

[0083] In the case of configuring the multi phases, at first, IC chipsare prepared by the number of the desired multi phases, and, from amongthe IC chips, one IC chip is selected as a master. Concretely, aselection signal SEL0 for selecting the master IC chip activates theoscillator OSC and a switch SWr, and 2 bits of selection signals SEL1and SEL2 specify the desired multi-phase number. Next, the master ICchip supplies the phase pulses φ2 to φ4, the reference voltage Vref, andthe outputs Vref+Δ and Vref−Δ from the upper and lower limit-voltagegenerating circuit VΔ. As a result, it turns out that φ2 to φ4, Vref,Vref+Δ, and Vref−Δ are added to the other IC chips, respectively. Thisallows the implementation of the multi-phasing.

[0084] Although, in the present embodiment, the multi-phase number hasbeen illustrated as 4, no limitation is imposed on the multi-phasenumber. The selection-signal number for setting the multi-phase numberis modified, and the circuit configuration of the phase shift circuitPSFT is modified to a circuit configuration that matches the multi-phasenumber, and these pieces of information are installed into the IC chips.This allows the multi-phase number to be increased or decreaseddepending on the requirements.

[0085] Embodiment 8

[0086]FIG. 17 illustrates an embodiment where the power-supply controlIC chip in the present invention is implemented on a printed wiringboard. In FIG. 17, the power-supply control ICs, and the inductor L andthe capacitor Co are mounted on a printed wiring board PB with the useof a BGA (: Ball Grid Array) and chip components, respectively, therebyallowing the downsized high-density implementation. Here, the capacitorCo is the chip ceramic capacitor. Incidentally, although notillustrated, in addition to these components, the CR circuit of thecapacitor C3 and the resistor R3, the boost circuit, and the inputcapacitor are mounted on the printed wiring board PB with the use ofchip components in this embodiment. Also, other than the on-chipmounting by the BGA, the CSP (: Chip Size Package) mounting may also beemployed.

[0087] Furthermore, in the case of the multi-phase compatibility, otherthan the on-chip mounting of the plural power-supply control ICs, theMCM (: Multi Chip Module) mounting may also be employed. In addition tothese mountings, components divided onto 2 IC chips, such as a controlunit including the error amplifier, the oscillator PWM, and the like,and a driver unit where the Power MOSFETs are built-in, may also bemounted on the printed wiring board in much the same way.

[0088] As described above, according to the present embodiment, itbecomes possible to implement the elimination of a pin neck, anenhancement in the heat-dissipating capability, and the downsizing ofthe printed wiring board of the power-supply device.

[0089] Embodiment 9

[0090]FIG. 18 illustrates the present embodiment. FIG. 18 illustratesthe embodiment that results from applying the present invention to HDDs(: Hard Disk Drives). Each of the HDDs includes a magnetic storage disk,a magnetic head, a magnetic-disk rotating drive, a magnetic-head drive,a magnetic-head position controller, and an input/output signalcontroller. DC-DC converters DC-DC1 to DC-DCn, i.e., the power-supplydevices described in the first to the eigth embodiments, supply electricpower to these HDDs HDD1 to HDDn. As the DC-DC converters DC-DC1 toDC-DCn, i.e., the power-supply devices illustrated in FIG. 18, thesingle-phase power-supply devices or the multi-phase power-supplydevices are used, depending on the current capacities of the HDDS, i.e.the targets of the power supply.

[0091] Embodiment 10

[0092] Next, the explanation will be given below concerning anembodiment where the control scheme in the present invention is appliedto isolation type DC-DC converters. FIG. 19 illustrates the embodimentapplied to a forward type converter. In FIG. 19, as is the case withFIG. 3, the CR smoothing filter of C and R is set up at both ends of aninductor L of the forward type converter. Next, the error amplifier EAgenerates an amplified error voltage, using the relationship between avoltage VFB at the midpoint of the CR smoothing filter and a referencevoltage Vref. Moreover, the use of the pulse-width modulation oscillatorPWM converts this amplified error voltage into a PWM pulse. This PWMpulse is passed through a transformer T2, and is applied to the gate ofa Power MOSFET QD for driving a transformer T1, then being subjected tothe negative-feedback control. This allows a desired output voltage tobe acquired in a stationary manner at the output terminal Vo. Thepresent method performs no negative-feedback control over the outputfrom the power LC filter, thereby making it possible to configure a highloop-stability power-supply system. Consequently, the present embodimentis especially effective when the ceramic capacitor is used as C of theLC filter.

[0093] Although, so far, the explanation has been given using the CRfilter in FIG. 3, the explanation is also possible using the method inFIG. 1. Also, instead of the transformer T2, the implementation is alsopossible using a photo coupler. In FIG. 19, the explanation has beengiven selecting the 1-stone forward type converter. The above-describedcontrol scheme, however, is also applicable to the other isolation typeDC-DC converters such as 2-stone forward type, push-pull type,half-bridge type, and full-bridge type.

[0094] Embodiment 11

[0095] Next, the illustration will be given below regarding anembodiment where the control scheme in the present invention is appliedto a commercially-available power-supply IC. FIG. 21 illustrates thecase where a PWM control IC HIP6311A and a driver-built-in Power MOSFETIC ILS6571 of the Intersil Corporation are used as thecommercially-available power-supply IC. The midpoint of C and R ofone-side CR smoothing filter set up at both ends of an inductor L isconnected to a feedback terminal FB of the PWM control IC HIP6311A. Themidpoint of C3 and R3 of the other-side CR smoothing filter is connectedto a transient variation detecting circuit TVD comprised of a referencepower-source LT1790A and a converter LT1715 of the Linear TechnologyCorporation through a high-input-impedance buffer amplifier BA and aresistor RN. Moreover, from the relationship between logical levels “H”and “L” of two signals a and b acquired by the transient variationdetecting circuit TVD, 3 operation state modes, i.e., a PWM pulse signalPWM1 (desired duty α ) outputted from the PWM control IC, a duty 0% α0,and a duty 100% α100, are switched selectively as indicated in FIG. 22by a selector HD74HC153, HD74HC157. Furthermore, its selected signal Yis outputted to a PWM terminal of the driver-built-in Power MOSFET IC.This shows that the control scheme in the present invention is alsoapplicable easily to the power-supply device configured using thecommercially-available power-supply IC. The application of the presentinvention is not limited to the products described in theabove-described embodiment. Incidentally, when the transient variationdetecting circuit TVD is not used, the PWM pulse signal PWM1 outputtedfrom the PWM control IC is directly connected to the PWM terminal of thedriver-built-in Power MOSFET IC, thereby making it possible to implementthe present invention.

[0096] It is needless to say that, although not illustrated, thepower-supply devices in the first to the eighth embodiments can beapplied and expanded to the other apparatuses, e.g., a VRM, a DC-DCconverter for portable appliances, and a general-purpose DC-DCconverter.

[0097] In the power-supply device of the present invention, none of thesecondary delay by the power LC smoothing filter enters the controlloop, which enhances the stability of the control loop. This furthermakes it possible to use the small-ESR chip ceramic capacitor in the LCsmoothing filter, thereby implementing the downsizing of thepower-supply device.

[0098] In the power-supply device of the present invention, the upperand lower limit value detecting circuit controls the high-speed responseat the transient state. This allows the power-supply device to makeresponse to even the high current slew rate (i.e., di/dt).

[0099] The power-supply device of the present invention can be easilymulti-phased. This makes it possible to simultaneously implement boththe large output current and the ripple-voltage reduction.

[0100] It should be further understood by those skilled in the art thatalthough the foregoing description has been made on embodiments of theinvention, the invention is not limited thereto and various changes andmodifications may be made without departing from the spirit of theinvention and the scope of the appended claims.

What is claimed is:
 1. A power-supply device including a step-down DC-DCconverter, comprising: power semiconductor switching components, drivingmeans for driving said power semiconductor switching components, apulse-width modulation oscillator for supplying said driving means witha driving signal, and an error amplifier for supplying said pulse-widthmodulation oscillator with an error signal indicating a comparisonresult between a reference value and an output power, wherein a controlloop of said power-supply device includes a power output filter throughwhich said output power passes, and a filter provided independently ofsaid power output filter, an output signal corresponding to said outputpower being fed back to said error amplifier after having passed throughsaid independently provided filter.
 2. The power-supply device accordingto claim 1, wherein said power output filter is an LC filter consistingof an inductor and a capacitor, said independently provided filter beinga CR filter consisting of a capacitor and a resistor, said CR filterbeing connected to said LC filter in parallel, and said output signalbeing fed back to said error amplifier after having passed through saidCR filter.
 3. The power-supply device according to claim 1, wherein saidpower output filter is an LC filter consisting of an inductor and acapacitor, said independently provided filter being a CR filterconsisting of a capacitor and a resistor, said CR filter being connectedto both ends of said inductor of said power output filter, and saidoutput signal being fed back to said error amplifier after having passedthrough said CR filter.
 4. The power-supply device according to claim 2[either claim 2 or claim 3], further comprising a transient variationdetecting circuit, said transient variation detecting circuit detectingsaid output power from an output terminal of said power output filter,and, if said output power has been found to exceed a predeterminedupper-limit value, outputting a signal for setting the duty α of saidpulse-width modulation oscillator at 0%, and, if said output power hasbeen found to be lower than a predetermined lower-limit value,outputting a signal for setting said duty a of said pulse-widthmodulation oscillator at 100%.
 5. The power-supply device according toclaim 2 [either claim 2 or claim 3], further comprising a transientvariation detecting circuit, said transient variation detecting circuitdetecting said output power from an output terminal of said CR filternewly provided at both ends of said inductor of said power outputfilter, and, if said output power has been found to exceed apredetermined upper-limit value, outputting a signal for setting theduty α of said pulse-width modulation oscillator at 0%, and, if saidoutput power has been found to be lower than a predetermined lower-limitvalue, outputting a signal for setting said duty α of said pulse-widthmodulation oscillator at 100%.
 6. A power-supply device of a step-downtype DC-DC converter including power-supply device units in pluralnumber, each of said plural power-supply device units, comprising: powersemiconductor switching components, driving means for driving said powersemiconductor switching components, a pulse-width modulation oscillatorfor supplying said driving means with a driving signal, and an erroramplifier for supplying said oscillator with an error signal of anoutput power, wherein each of said plural power-supply device unitsincludes a power output filter through which said output power passes,and a filter provided independently of said power output filter, anoutput signal being fed back to said error amplifier after having passedthrough said independently provided filter.
 7. The power-supply deviceaccording to claim 6, wherein, in order to perform a parallel operationof said plural power-supply device units, said plural power-supplydevice units include said pulse-width modulation oscillators in commontherewith, phases of said output driving signals from said pulse-widthmodulation oscillators being shifted, and said signals of which phaseshave been shifted being supplied to said plural power-supply deviceunits.
 8. The power-supply device according to claim 7, wherein each ofsaid plural power-supply device units further comprises a transientvariation detecting circuit, said transient variation detecting circuitdetecting said output power from an output terminal of said power outputfilter, and, if said output power has been found to exceed apredetermined upper-limit value, outputting a signal for setting theduty α of said pulse-width modulation oscillator at 0%, and, if saidoutput power has been found to be lower than a predetermined lower-limitvalue, outputting a signal for setting said duty α of said pulse-widthmodulation oscillator at 100%.
 9. The power-supply device according toclaim 1, wherein said power semiconductor switching components, saiddriving means for driving said power semiconductor switching components,said pulse-width modulation oscillator, said error amplifier, and atransient variation detecting circuit are formed on one and the samesemiconductor board, said transient variation detecting circuitdetecting said output power from an output terminal of said power outputfilter, and, if said output power has been found to exceed apredetermined upper-limit value, outputting a signal for setting theduty α of said pulse-width modulation oscillator at 0%, and, if saidoutput power has been found to be lower than a predetermined lower-limitvalue, outputting a signal for setting said duty α of said pulse-widthmodulation oscillator at 100%.
 10. A hard-disk device which includes amagnetic storage disk, a magnetic head, a magnetic-disk rotating drive,a magnetic-head drive, a magnetic-head position controller, aninput/output signal controller, and a power-supply device for supplyingpower, said power-supply device being a step-down type DC-DC converterwhich comprises power semiconductor switching components, driving meansfor driving said power semiconductor switching components, a pulse-widthmodulation oscillator for supplying said driving means with a drivingsignal, and an error amplifier for supplying said oscillator with anerror signal of an output power, wherein said step-down type DC-DCconverter further comprises a power output filter through which saidoutput power passes, and a filter provided independently of said poweroutput filter, an output signal being fed back to said error amplifierafter having passed through said independently provided filter, andwherein said power output filter is an LC filter consisting of aninductor and a capacitor, said independently provided filter being a CRfilter consisting of a capacitor and a resistor, said CR filter beingconnected to said LC filter in parallel, and said output signal beingfed back to said error amplifier after having passed through said CRfilter.
 11. A power-supply device implemented by applying saidpower-supply device according to claim 1 to an isolation type DC-DCconverter.
 12. A power-supply device implemented by using a combinationof an oscillator, a one-shot multivibrator, and a voltage-to-currentconverter as said pulse-width modulation oscillator of said power-supplydevice according to claim
 1. 13. A multi-phase control power-supplydevice, comprising: a phase shift circuit, and said pulse-widthmodulation oscillator of said power-supply device according to claim 12,wherein said phase shift circuit is provided after said oscillator, saidone-shot multivibrator being provided in correspondence with each phase.14. A power-supply device according to claim 1, wherein said erroramplifier has a low input impedance, and is connected to the output ofsaid independently provided filter through a buffer amplifier having ahigh input impedance.
 15. An integrated circuit wherein a power-supplydevice according to claim 1 is built in a semiconductor chip.